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AD8339 数据手册DataSheet 下载
DC to 50 MHz, Quad I/Q Demodulator
and Phase Shifter
AD8339
FEATURES
APPLICATIONS
Medical imaging (CW ultrasound beamforming)
Phased array systems
Radar
Adaptive antennas
Communication receivers
FUNCTIONAL BLOCK DIAGRAM
RF1N
RF1P
AD8339
Φ
I1OP
Φ
Q1OP
Φ
I2OP
Φ
Q2OP
Φ
I3OP
Φ
Q3OP
Φ
I4OP
Φ
Q4OP
SCLK
SDI
SDO
SERIAL
INTERFACE
CSB
RF2P
RF2N
4LOP
4LON
0°
÷4
90°
RF3P
RF3N
VPOS
BIAS
VNEG
RF4N
RF4P
www.BDTIC.com/ADI
06587-001
Quad integrated I/Q demodulator
16 phase select on each output (22.5° per step)
Quadrature demodulation accuracy
Phase accuracy: ±1°
Amplitude balance: ±0.25 dB
Bandwidth
4LO: LF to 200 MHz
RF: LF to 50 MHz
Baseband: determined by external filtering
Output dynamic range: 160 dB/Hz
LO drive: >0 dBm (50 Ω), single-ended sine wave
Supply: ±5 V
Power consumption: 73 mW/channel (290 mW total)
Power-down via SPI (each channel and complete chip)
Figure 1.
GENERAL DESCRIPTION
The AD8339 1 is a quad I/Q demodulator configured to be
driven by a low noise preamplifier with differential outputs. It is
optimized for the LNA in the AD8332/AD8334/AD8335 family
of VGAs. The part consists of four identical I/Q demodulators
with a 4× local oscillator (LO) input that divides this signal and
generates the necessary 0° and 90° phases of the internal LO
that drive the mixers. The four I/Q demodulators can be used
independently of each other (assuming that a common LO is
acceptable) because each has a separate RF input.
Continuous wave (CW) analog beamforming (ABF) and I/Q
demodulation are combined in a single 40-lead ultracompact
chip scale device, making the AD8339 particularly applicable in
high density ultrasound scanners. In an ABF system, time
domain coherency is achieved following the appropriate phase
alignment and summation of multiple receiver channels. A reset
pin synchronizes multiple ICs to start each LO divider in the
same quadrant. Sixteen programmable 22.5° phase increments
are available for each channel. For example, if Channel 1 is used
as a reference and Channel 2 has an I/Q phase lead of 45°, then
the user can phase align Channel 2 with Channel 1 by choosing
the correct code.
1
The mixer outputs are in current form for convenient summation. The independent I and Q mixer output currents are summed
and converted to a voltage by a low noise, high dynamic range,
current-to-voltage (I-V) transimpedance amplifier, such as the
AD8021 or the AD829. Following the current summation, the
combined signal is applied to a high resolution analog-to-digital
converter (ADC), such as the AD7665 (16-bit, 570 kSPS).
An SPI-compatible serial interface port is provided to easily
program the phase of each channel; the interface allows daisy
chaining by shifting the data through each chip from SDI to SDO.
The SPI also allows for power-down of each individual channel
and the complete chip. During power-down, the serial interface
remains active so that the device can be programmed again.
The dynamic range is typically 160 dB/Hz at the I and Q
outputs. The AD8339 is available in a 6 mm × 6 mm, 40-lead
LFCSP and is specified over the industrial temperature range of
−40°C to +85°C.
Patent pending.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113
©2007 Analog Devices, Inc. All rights reserved.
AD8339
TABLE OF CONTENTS
Features .............................................................................................. 1
Summation of Multiple Channels (Analog Beamforming).. 20
Applications....................................................................................... 1
Phase Compensation and Analog Beamforming................... 21
Functional Block Diagram .............................................................. 1
Channel Summing ..................................................................... 21
General Description ......................................................................... 1
Serial Interface ................................................................................ 22
Revision History ............................................................................... 2
ENBL Bits .................................................................................... 22
Specifications..................................................................................... 3
Applications..................................................................................... 23
Absolute Maximum Ratings............................................................ 5
Logic Inputs and Interfaces....................................................... 23
ESD Caution.................................................................................. 5
Reset Input .................................................................................. 23
Pin Configuration and Function Descriptions............................. 6
LO Input ...................................................................................... 23
Equivalent Input Circuits ................................................................ 7
Evaluation Board ............................................................................ 24
Typical Performance Characteristics ............................................. 8
Connections to the Board ......................................................... 25
Test Circuits..................................................................................... 14
Test Configurations.................................................................... 25
Theory of Operation ...................................................................... 18
AD8339-EVALZ Artwork ......................................................... 32
Quadrature Generation ............................................................. 19
Outline Dimensions ....................................................................... 36
I/Q Demodulator and Phase Shifter ........................................ 19
Ordering Guide .......................................................................... 36
Dynamic Range and Noise........................................................ 19
www.BDTIC.com/ADI
REVISION HISTORY
8/07—Revision 0: Initial Version
Rev. 0 | Page 2 of 36
AD8339
SPECIFICATIONS
VS = ±5 V, TA = 25°C, 4fLO = 20 MHz, fRF = 5.01 MHz, fBB = 10 kHz, PLO ≥ 0 dBm, per channel performance, dBm (50 Ω), unless otherwise
noted. Single-channel AD8021 LPF values: RFILT = 787 Ω and CFILT = 2.2 nF (see Figure 52).
Table 1.
Parameter
OPERATING CONDITIONS
Local Oscillator Frequency Range
RF Frequency Range
Baseband Bandwidth
LO Input Level
VSUPPLY (VS)
Temperature Range
DEMODULATOR PERFORMANCE
Input Impedance
Transconductance
Dynamic Range
Maximum Input Swing
Peak Output Current (No Filtering)
Conditions
Min
4× internal LO at Pin 4LOP and Pin 4LON, square wave
drive via LVDS (see Figure 62)
Mixing
Limited by external filtering
Max
Unit
0.01
200
MHz
DC
DC
50
50
13
±5.5
+85
MHz
MHz
dBm
V
°C
±4.5
−40
Typ
0
±5
RF, differential
LO, differential
Demodulated IOUT/VIN; each Ix or Qx output after lowpass filtering measured from RF inputs, all phases
IP1dB − input referred noise (dBm)
25||10
100||4
1.15
kΩ||pF
kΩ||pF
mS
160
Differential; inputs biased at 2.5 V; Pin RFxP, Pin RFxN
0° phase shift
45° phase shift
Ref = 50 Ω
Ref = 1 V rms
fRF1 = 5.010 MHz, fRF2 = 5.015 MHz, fLO = 5.023 MHz
Baseband tones: 0 dBm @ 8 kHz and 13 kHz
Baseband tones: −1 dBm @ 8 kHz and −31 dBm @ 13 kHz
fRF1 = 5.010 MHz, fRF2 = 5.015 MHz, fLO = 5.023 MHz
Measured at RF inputs, worst phase, measured into 50 Ω
Measured at baseband outputs, worst phase, AD8021
disabled, measured into 50 Ω
All codes, see Figure 41
Output noise/conversion gain (see Figure 46)
Output noise/RFILT
With AD8334 LNA
RS = 50 Ω, RFB = ∞
RS = 50 Ω, RFB = 1.1 kΩ
RS = 50 Ω, RFB = 274 Ω
Pin 4LOP and Pin 4LON
Pin RFxP and Pin RFxN
Pin 4LOP and Pin 4LON (each pin)
For maximum differential swing; Pin RFxP and Pin RFxN
(dc-coupled to AD8334 LNA output)
Pin IxOP and Pin QxOP
One channel is reference, others are stepped
16 phase steps per channel
Ix to Qx; all phases, 1σ
Ix to Qx; all phases, 1σ
Phase match I-to-I and Q-to-Q; −40°C < TA < +85°C
Amplitude match I-to-I and Q-to-Q; −40°C < TA < +85°C
2.8
±2.4
±3.1
14.8
1.85
dB (1 Hz
BW)
V p-p
mA
mA
dBm
dBV
−60
−66
31
−118
−68
dBc
dBc
dBm
dBm
dBm
−1.3
11.8
12.9
dB
nV/√Hz
pA/√Hz
8.4
9.1
11.5
−3
−45
dB
dB
dB
μA
μA
V
V
www.BDTIC.com/ADI
Input P1dB
Third-Order Intermodulation (IM3)
Equal Input Levels
Unequal Input Levels
Third-Order Input Intercept (IIP3)
LO Leakage
Conversion Gain
Input Referred Noise
Output Current Noise
Noise Figure
Bias Current
LO Common-Mode Range
RF Common-Mode Voltage
Output Compliance Range
PHASE ROTATION PERFORMANCE
Phase Increment
Quadrature Phase Error
I/Q Amplitude Imbalance
Channel-to-Channel Matching
Rev. 0 | Page 3 of 36
0.2
3.8
2.5
−1.5
−2
+0.7
22.5
±1
±0.05
±1
±0.1
+2
V
Degrees
Degrees
dB
Degrees
dB
AD8339
Parameter
LOGIC INTERFACES
Logic Level High
Logic Level Low
Logic Level High
Logic Level Low
Bias Current
Input Resistance
LO Divider RSET Setup Time
LO Divider RSET High Pulse Width
LO Divider RSET Response Time
Phase Response Time
Enable Response Time
Output
Logic Level High
Logic Level Low
SPI TIMING CHARACTERISTICS
SCLK Frequency
CSB Fall to SCLK Setup Time
SCLK High Pulse Width
SCLK Low Pulse Width
Data Access Time (SDO) After SCLK
Rising Edge
Data Setup Time Before SCLK Rising
Edge
Data Hold Time After SCLK
Rising Edge
SCLK Rise to CSB Rise Hold Time
CSB Rise to SCLK Rise Hold Time
POWER SUPPLY
Supply Voltage
Current
Conditions
Min
Pin SDI, Pin CSB, Pin SCLK, Pin RSET
1.5
Typ
Max
0.9
Pin RSTS
1.8
1.2
Logic high (pulled to +5 V)
Logic low (pulled to GND)
RSET rising or falling edge to 4LOP to 4LON (differential)
rising edge
0.5
0
4
5
20
200
5
12
Measured from CSB going high
Measured from CSB going high (with 0.1 μF capacitor on
Pin LODC); no channel enabled
At least one channel enabled
Pin SDO loaded with 5 pF and next SDI input
500
1.7
Pin SDI, Pin SDO, Pin CSB, Pin SCLK, Pin RSTS
fCLK
t1
t2
t3
t4
15
1.9
0.2
Quiescent Power
Disable Current
PSRR
V
V
V
V
μA
μA
MΩ
ns
ns
ns
μs
μs
ns
0.5
10
0
10
10
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Over Temperature
−40°C < TA < +85°C
Unit
100
V
V
MHz
ns
ns
ns
ns
t5
2
ns
t6
2
ns
t7
t8
Pin VPOS, Pin VNEG
15
0
ns
ns
±4.5
VPOS, all phase bits = 0
VNEG, all phase bits = 0
VPOS, all phase bits = 0
VNEG, all phase bits = 0
Per channel, all phase bits = 0
Per channel maximum (depends on phase bits)
All channels disabled; SPI stays on
VPOS to Ix/Qx outputs, @ 10 kHz
VNEG to Ix/Qx outputs, @ 10 kHz
Rev. 0 | Page 4 of 36
±5
35
−18
33
±5.5
36
−19
−17
66
88
2.75
−85
−85
V
mA
mA
mA
mA
mW
mW
mA
dB
dB
AD8339
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter
Voltages
Supply Voltage (VS)
RF Inputs
4LO Inputs
Outputs (IxOP, QxOP)
Digital Inputs
SDO Output
LODC Pin
Thermal Data (4-Layer JEDEC Board,
No Air Flow, Exposed Pad Soldered
to PC Board)
θJA
θJB
θJC
ψJT
ψJB
Maximum Junction Temperature
Maximum Power Dissipation
(Exposed Pad Soldered to PC Board)
Operating Temperature Range
Storage Temperature Range
Lead Temperature (Soldering, 60 sec)
Rating
±6 V
+6 V to GND
+6 V to GND
+0.7 V to −6 V
+6 V to GND
+6 V to GND
VPOS − 1.5 V to +6 V
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
ESD CAUTION
32.2°C/W
17.8°C/W
2.7°C/W
0.3°C/W
16.7°C/W
150°C
2W
−40°C to +85°C
−65°C to +150°C
300°C
www.BDTIC.com/ADI
Rev. 0 | Page 5 of 36
AD8339
VNEG
Q1OP
I1OP
VPOS
RSET
COMM
RF1N
RF1P
RSTS
SDI
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
40 39 38 37 36 35 34 33 32 31
RF2P
1
2
COMM
3
COMM
SCLK
4
5
CSB
6
VPOS
7
VPOS
RF3P
RF2N
RF3N
PIN 1
INDICATOR
AD8339
30
Q2OP
29
28
27
26
I2OP
VPOS
VPOS
25
24
4LOP
4LON
VNEG
8
23
VNEG
9
10
22
I3OP
21
Q3OP
TOP VIEW
(Not to Scale)
06587-002
VNEG
I4OP
Q4OP
VPOS
LODC
COMM
RF4P
RF4N
VPOS
SDO
11 12 13 14 15 16 17 18 19 20
Figure 2. 40-Lead LFCSP
Table 3. Pin Function Descriptions
Pin No.
1, 2, 9, 10, 13,
14, 37, 38
3, 4, 15, 36
5
6
7, 8, 11, 16,
27, 28, 35
Mnemonic
RF1P to RF4P,
RF1N to RF4N
COMM
SCLK
CSB
VPOS
Description
RF Inputs. Require external 2.5 V bias for optimum symmetrical input differential swing if ±5 V supplies
are used.
Ground.
Serial Interface—Clock.
Serial Interface—Chip Select Bar. Active low.
Positive Supply. These pins should be decoupled with a ferrite bead in series with the supply, plus a
0.1 μF and 1 nF capacitor between the VPOS pins and ground. Because the VPOS pins are internally
connected, one set of supply decoupling components on each side of the chip should be sufficient.
Serial Interface—Data Output. Normally connected to SDI of next chip or left open.
Decoupling Pin for LO. A 0.1 μF capacitor should be connected between this pin and ground. Value of
capacitor influences chip enable/disable times.
I/Q Outputs. These outputs provide a bidirectional current that can be converted back to a voltage via a
transimpedance amplifier. Multiple outputs can be summed together by simply connecting them (wireOR). The bias voltage should be set to 0 V or less by the transimpedance amplifier (see Figure 52).
Negative Supply. These pins should be decoupled with a ferrite bead in series with the supply, plus a
0.1 μF and 1 nF capacitor between the pin and ground. Because the VNEG pins are internally connected,
one set of supply decoupling components should be sufficient.
LO Inputs. No internal bias; optimally biased by an LVDS driver. For best performance, these inputs
should be driven differentially. If driven by single-ended sine wave at 4LOP or 4LON, the signal level
should be > 0 dBm (50 Ω) with external bias resistors.
LO Interface—Reset. Logic threshold is at ~1.3 V and therefore can be driven by >1.8 V CMOS logic.
Serial Interface—Data Input. Logic threshold is at ~1.3 V and therefore can be driven by >1.8 V CMOS
logic.
Reset for SPI Interface. Logic threshold is at ~1.5 V with ± 0.3 V hysteresis and should be driven by >3.3 V
CMOS logic. For quick testing without the need to program the SPI, the voltage on the RSTS pin should
be pulled to −1.4 V; this enables all four channels in the phase (I = 1, Q = 0) state.
www.BDTIC.com/ADI
12
17
SDO
LODC
18, 19, 21, 22,
29, 30, 32, 33
I1OP to I4OP,
Q1OP to Q4OP
20, 23, 24, 31
VNEG
25, 26
4LOP, 4LON
34
39
RSET
SDI
40
RSTS
Rev. 0 | Page 6 of 36
AD8339
EQUIVALENT INPUT CIRCUITS
VPOS
VPOS
SCLK
CSB
SDI
RSET
RSTS
RFxP
LOGIC
INTERFACE
06587-006
COMM
06587-003
RFxN
COMM
Figure 3. Logic Inputs
Figure 6. RF Inputs
VPOS
COMM
4LOP
IxOP
QxOP
4LON
COMM
VNEG
Figure 7. Output Drivers
Figure 4. Local Oscillator Inputs
VPOS
COMM
06587-005
LODC
Figure 5. LO Decoupling Pin
Rev. 0 | Page 7 of 36
06587-007
06587-004
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AD8339
TYPICAL PERFORMANCE CHARACTERISTICS
VS = ±5 V, TA = 25°C, 4fLO = 20 MHz, fLO = 5 MHz, fRF = 5.01 MHz, fBB = 10 kHz, 4fLO − LVDS drive; per channel performance shown is
typical of all channels, differential voltages, dBm (50 Ω), phase select code = 0000, unless otherwise noted (see default test circuit).
2
CODE 0100
CODE 0011
IMAGINARY (Normalized)
1.0
CODE 0010
CODE 0001
Q
0.5
CODE 1000
CODE 0000
0
I
–0.5
CODE 1100
–1.5
–1.5
–1.0
–0.5
0
–1
0
0.5
1.0
1.5
CHANNEL 3
CHANNEL 4
–2
2
F = 1MHz
1
0
06587-008
–1.0
–2.0
F = 5MHz
1
–1
CHANNEL 3
CHANNEL 4
–2
0010
0100
0000
2.0
REAL (Normalized)
06587-011
f = 1MHz
AMPLITUDE ERROR (Degrees)
1.5
0110
1000
1010
1100
1110
1111
CODE (Binary)
Figure 8. Normalized Vector Plot of Phase, CH2, CH3, and CH4 vs. CH1, CH1 Is
Fixed at 0°, CH2, CH3, and CH4 Stepped 22.5°/Step, All Codes Displayed
Figure 11. Representative Phase Error vs. Binary Phase-Select Code
at 1 MHz and 5 MHz, CH3 and CH4 Are Displayed With Respect to CH1
360
315
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PHASE (Degrees)
270
5MHz
225
1
2
180
1MHz
135
0
0000
06587-009
45
06587-012
90
0010
0100
0110
1000
1010
1100
1110
C2 500mV Ω
C4 500mV Ω
1111
CODE (Binary)
Figure 9. Representative Phase Delay vs. Binary Phase-Select Code
at 1 MHz and 5 MHz, CH3 and CH4 Are Displayed With Respect to CH1
R1 500mV 20µs
R2 500mV 20µs
20.0µs/DIV
2.5MS/s 400ns/PT
A C2
30.0mV
Figure 12. Representative Phase Delays of the I or Q Outputs,
CH2 Is Displayed With Respect to CH1, for Delays of 22.5°, 45°, 67.5°, and 90°
1
1.0
I OUTPUT OF CHANNEL 1 SHOWN
F = 5MHz
0.5
CHANNEL 3
CHANNEL 4
–1.0
1.0
F = 1MHz
0.5
0
CHANNEL 3
CHANNEL 4
–1.0
0000
0010
0100
0110
1000
1010
1100
1110
–1
–2
CODE 0000
CODE 0001
CODE 0010
CODE 0011
06587-010
–0.5
0
–3
1M
1111
06587-013
–0.5
CONVERSION GAIN (dB)
AMPLITUDE ERROR (dB)
0
10M
50M
RF FREQUENCY (Hz)
CODE (Binary)
Figure 10. Representative Amplitude Error vs. Binary Phase-Select Code
at 1 MHz and 5 MHz, CH3 and CH4 Are Displayed With Respect to CH1
Rev. 0 | Page 8 of 36
Figure 13. Conversion Gain vs. RF Frequency, First Quadrant,
Baseband Frequency = 10 kHz
AD8339
0.5
0.4
4
2
0
–2
–4
–6
–8
1M
10M
0.3
0.2
0.1
0
–0.1
–0.2
–0.3
06587-017
I/Q AMPLITUDE IMBALANCE (dB)
6
06587-014
QUADRATURE PHASE ERROR (Degrees)
8
–0.4
–0.5
100
50M
1k
RF FREQUENCY (Hz)
10k
100k
BASEBAND FREQUENCY (Hz)
Figure 14. Representative Range of Quadrature Phase Error vs. RF Frequency
for All Channels and Codes
Figure 17. Representative Range of I/Q Amplitude Imbalance vs. Baseband
Frequency for All Channels and Codes (See Figure 43)
2.0
3
2
AMPLITUDE ERROR (dB)
1.0
0.5
0
–1.0
–1.5
1
0
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–2.0
100
–1
–2
1k
10k
–3
1M
100k
10M
BASEBAND FREQUENCY (Hz)
50M
RF FREQUENCY (Hz)
Figure 15. Representative Range of Quadrature Phase Error vs. Baseband
Frequency for All Channels and Codes (See Figure 43)
Figure 18. Typical Channel-to-Channel Amplitude Match vs. RF Frequency,
First Quadrant, Over the Range of Operating Temperatures
0.5
8
fBB = 10kHz
0.4
6
PHASE ERROR (Degrees)
0.3
0.2
0.1
0
–0.1
–0.2
4
2
0
–2
–4
–0.3
10M
–8
1M
50M
RF FREQUENCY (Hz)
06587-019
–0.4
–0.5
1M
–6
06587-016
I/Q AMPLITUDE IMBALANCE (dB)
06587-018
–0.5
06587-015
QUADRATURE PHASE ERROR (Degrees)
fBB = 10kHz
1.5
10M
50M
RF FREQUENCY (Hz)
Figure 16. Representative Range of I/Q Amplitude Imbalance vs.
RF Frequency for All Channels and Codes
Figure 19. Typical Channel-to-Channel Phase Error vs. RF Frequency,
First Quadrant, Over the Range of Operating Temperatures
Rev. 0 | Page 9 of 36
AD8339
1.4
0
I OUTPUT OF CHANNEL 1 SHOWN
TRANSCONDUCTANCE = [(VBB/787Ω)/V RF]
–10
–20
IM3 (dBc)
1.2
1.1
–30
3
8 13 18
IM3 PRODUCTS
LO = 5.023MHz
RF1 = 5.015MHz
RF2 = 5.010MHz
–40
–50
–60
0.8
1M
10M
–70
1M
50M
RF FREQUENCY (Hz)
10M
50M
RF FREQUENCY (Hz)
Figure 23. Representative Range of IM3 vs. RF Frequency, First Quadrant
(See Figure 48)
Figure 20. Transconductance vs. RF Frequency
for First Quadrant Phase Delays
10
35
+85°C
+25°C
–40°C
0
30
–10
25
OIP3 (dBm)
–20
–30
20
15
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–40
10
5
06587-021
–60
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
0
1M
5.0
COMMON MODE VOLTAGE (V)
10M
06587-024
–50
–70
06587-023
PHASE DELAY = 0°
PHASE DELAY = 22.5°
PHASE DELAY = 45°
PHASE DELAY = 67.5°
0.9
CONVERSION GAIN (dB)
0dBm
1.0
06587-020
TRANSCONDUCTANCE (mS)
1.3
50M
RF FREQUENCY (Hz)
Figure 24. Representative Range of OIP3 vs. RF Frequency, First Quadrant
(See Figure 48)
Figure 21. LO Common-Mode Range at Three Temperatures
20
35
18
30
16
25
OIP3 (dBm)
12
10
8
6
20
15
10
4
10M
0
1k
50M
RF FREQUENCY (Hz)
06587-025
2
0
1M
5
06587-022
IP1dB (dBm)
14
10k
100k
BASEBAND FREQUENCY (Hz)
Figure 22. Representative Range of IP1dB vs. RF Frequency,
Baseband Frequency = 10 kHz, First Quadrant (See Figure 42)
Figure 25. Representative Range of OIP3 vs. Baseband Frequency
(See Figure 47)
Rev. 0 | Page 10 of 36
AD8339
0
20
LO LEVEL = 0dBm
18
–10
16
NOISE FIGURE (dB)
–30
–40
–50
–60
–70
14
12
10
8
6
4
06587-026
–80
–90
1M
10M
06587-029
LO LEAKAGE (dBm)
–20
2
0
1M
50M
10M
RF FREQUENCY (Hz)
Figure 29. Noise Figure vs. RF Frequency (When Driven by AD8334 LNA)
Figure 26. Representative Range of LO Leakage vs. RF Frequency
at I and Q Outputs
0
50M
RF FREQUENCY (Hz)
172
LO LEVEL = 0dBm
170
–20
DYNAMIC RANGE (dB)
–60
–100
164
162
www.BDTIC.com/ADI
160
Q1
Q2
Q3
Q4
I1 + I2
I3 + I4
Q1 + Q2
Q3 + Q4
I1 + I2 + I3 + I4
Q1 + Q2 + Q3 + Q4
158
–120
06587-027
156
–140
1M
10M
154
152
1M
50M
10M
RF FREQUENCY (Hz)
0
14
–144.1
–2
12
–145.4
–4
10
–147.0
–6
8
–148.9
6
–151.4
4
–154.9
–12
2
–161.0
–14
GAIN (dB)
NOISE (dBm)
–142.9
GAIN = VBB/VRF
–8
–10
06587-028
NOISE (nV/√Hz)
Figure 30. Dynamic Range vs. RF Frequency, IP1dB Minus Noise Level
16
10M
50M
RF FREQUENCY (Hz)
Figure 27. Representative Range of LO Leakage vs. RF Frequency at RF Inputs
0
1M
06587-030
–80
166
–16
–3.5
50M
RF FREQUENCY (Hz)
DELAY = 0°
DELAY = 22.5°
DELAY = 45°
DELAY = 67.5°
–3.0
–2.5
–2.0
–1.5
–1.0
–0.5
06587-031
LO LEAKAGE (dBm)
168
–40
0
0.5
1.0
VOLTAGE (V)
Figure 28. Representative Range of Input Referred Noise vs. RF Frequency
Figure 31. Output Compliance Range for Four Values of Phase Delay
(See Figure 49)
Rev. 0 | Page 11 of 36
AD8339
T
CH3 AMPL
3.18V
CH3 AMPL
5.04 V
3
3
CH2 AMPL
790mV
CH2 AMPL
370mV
2
CH2 500mV CH3 1.00V Ω
M200ns
A CH3
T
608.000ns
06587-035
06587-032
2
600mV
CH3 2.00V Ω CH2 500mV
Figure 32. Enable Response vs. CSB (Filter Disabled to Show Response)
with a Previously Enabled Channel (See Figure 43)
3
2.52mV
Figure 35. LO Reset Response (see Figure 44)
3
www.BDTIC.com/ADI
CH2 AMPL
1.82V
CH3 1.00V Ω CH2 500mV
M2.00µs
A CH3
T
7.840µs
06587-033
2
780mV
CH3 1.00V Ω CH2 1.00V
CH4 1.00V
Figure 33. Enable Response vs. CSB (Filter Disabled to Show Response) with
No Channels Previously Enabled (See Figure 43)
M40.0µs
A CH3
T
46.4000µs
06587-036
2
M200µs
A CH3
T
–175.200ns
640mV
Figure 36. Phase Switching Response at 45° (Top: CSB)
CH3 AMPL
3.18V
3
3
CH2 AMPL
210mV
06587-034
CH3 1.00V Ω CH2 500mV
M200µs
A CH3
T
–492.00ns
06587-037
2
2
600mV
CH3 1.00V Ω CH2 1.00V
CH4 1.00V
Figure 34. Disable Response vs. CSB (Top: CSB)
(See Figure 43)
M40.0µs
A CH3
T
46.4000µs
640mV
Figure 37. Phase Switching Response at 90° (Top: CSB)
Rev. 0 | Page 12 of 36
AD8339
60
SUPPLY CURRENT (mA)
50
3
2
40
VPOS
30
20
VNEG
CH3 1.00V Ω CH2 1.00V
CH4 1.00V
M40.0µs
A CH3
T
46.4000µs
0
–50
640mV
06587-040
06587-038
10
–30
–10
10
30
50
TEMPERATURE (°C)
Figure 38. Phase Switching Response at 180° (Top: CSB)
Figure 40. Supply Current vs. Temperature
0
–10
–20
–40
–50
–60
–70
–80
–90
VNEG
www.BDTIC.com/ADI
–100
10k
VPOS
06587-039
PSRR (dB)
–30
100k
1M
10M
50M
FREQUENCY (Hz)
Figure 39. PSRR vs. Frequency (see Figure 50)
Rev. 0 | Page 13 of 36
70
90
AD8339
TEST CIRCUITS
AD8021
120nH
FB
0.1µF
787Ω
AD8334
LNA
20Ω
LPF
50Ω
2.2nF
RFxP
Ix
AD8339
0.1µF
20Ω
RFxN
OSCILLOSCOPE
2.2nF
Qx
LOP
787Ω
SIGNAL
GENERATOR
50Ω
AD8021
06587-041
SIGNAL
GENERATOR
Figure 41. Default Test Circuit
AD8021
120nH
FB
0.1µF
100Ω
AD8334
LNA
20Ω
LPF
10nF
RFxP
Ix
AD8339
RFxN
OSCILLOSCOPE
10nF
Qx
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50Ω
0.1µF
20Ω
LOP
100Ω
SIGNAL
GENERATOR
50Ω
AD8021
06587-042
SIGNAL
GENERATOR
Figure 42. P1dB Test Circuit
AD8021
120nH
FB
0.1µF
AD8334
LNA
20Ω
LPF
50Ω
RFxP
Ix
787Ω
OSCILLOSCOPE
AD8339
0.1µF
20Ω
RFxN
Qx
787Ω
LOP
SIGNAL
GENERATOR
50Ω
AD8021
Figure 43. Phase and Amplitude vs. Baseband Frequency
Rev. 0 | Page 14 of 36
06587-043
SIGNAL
GENERATOR
AD8339
AD8021
120nH
FB
0.1µF
AD8334
LNA
20Ω
RFxP
LPF
787Ω
Ix
OSCILLOSCOPE
AD8339
50Ω
0.1µF
RFxN
RSET
20Ω
787Ω
Qx
LOP
SIGNAL
GENERATOR
50Ω
50Ω
SIGNAL
GENERATOR
06587-044
SIGNAL
GENERATOR
AD8021
Figure 44. LO Reset Response
120nH
FB
0.1µF
AD8334
LNA
OSCILLOSCOPE
20Ω
RFxP
LPF
50Ω
Ix
AD8339
0.1µF
RFxN
20Ω
50Ω 50Ω
Qx
LOP
SIGNAL
GENERATOR
50Ω
06587-045
SIGNAL
GENERATOR
www.BDTIC.com/ADI
Figure 45. RF Input Range
AD829
6.98kΩ
AD8334
LNA
270pF
20Ω
RFxP
0.1µF
0.1µF
20Ω
Ix
AD8339
RFxN
SPECTRUM
ANALYZER
270pF
Qx
LOP
6.98kΩ
50Ω
SIGNAL
GENERATOR
Figure 46. Noise
Rev. 0 | Page 15 of 36
AD829
06587-046
120nH
FB
0.1µF
AD8339
AD8021
SPLITTER
AD8334
–9.5dB
LNA
120nH
20Ω
FB
0.1µF
50Ω
787Ω
100pF
RFxP
Ix
AD8339
SIGNAL
GENERATOR
50Ω
0.1µF
RFxN
20Ω
SPECTRUM
ANALYZER
100pF
Qx
LOP
787Ω
SIGNAL
GENERATOR
50Ω
AD8021
06587-047
SIGNAL
GENERATOR
Figure 47. OIP3 vs. Baseband Frequency
AD8021
SPLITTER
AD8334
–9.5dB
LNA
120nH
20Ω
FB
0.1µF
50Ω
787Ω
2.2nF
RFxP
Ix
AD8339
SIGNAL
GENERATOR
50Ω
0.1µF
RFxN
20Ω
SPECTRUM
ANALYZER
2.2nF
Qx
LOP
787Ω
SIGNAL
GENERATOR
www.BDTIC.com/ADI
50Ω
AD8021
06587-048
SIGNAL
GENERATOR
Figure 48. OIP3 and IM3 vs. RF Frequency
AD8021
787Ω
AD8334
LNA
20Ω
LPF
50Ω
2.2nF
RFxP
Ix
AD8339
0.1µF
20Ω
RFxN
OSCILLOSCOPE
2.2nF
Qx
LOP
SIGNAL
GENERATOR
787Ω
50Ω
SIGNAL
GENERATOR
AD8021
Figure 49. Output Compliance Range
Rev. 0 | Page 16 of 36
06587-049
120nH
FB
0.1µF
AD8339
SIGNAL
GENERATOR
VPOS
VPOS
RFxP
0.1µF
Ix
SPECTRUM
ANALYZER
AD8339
RFxN
Qx
LOP
SIGNAL
GENERATOR
06587-050
VPOS
Figure 50. PSRR
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Rev. 0 | Page 17 of 36
AD8339
THEORY OF OPERATION
RF2N
1
RF2P
2
COMM
3
COMM
4
SCLK
5
RSTS
SDI
RF1P
RF1N
40
39
38
37
COMM VPOS
36
RSET
I1OP
Q1OP
VNEG
34
33
32
31
35
0°
Φ
CURRENT
MIRROR
30
Q2OP
Φ
CURRENT
MIRROR
29
I2OP
Φ
CURRENT
MIRROR
28
VPOS
Φ
CURRENT
MIRROR
27
VPOS
26
4LOP
25
4LON
V TO I
V TO I
SERIAL
INTERFACE
(SPI)
0°
LOCAL OSCILLATOR DIVIDE BY 4
90°
CSB
6
VPOS
7
Φ
CURRENT
MIRROR
24
VNEG
Φ
CURRENT
MIRROR
23
VNEG
V TO I
VPOS
8
BIAS
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RF3P
9
Φ
CURRENT
MIRROR
22
I3OP
Φ
CURRENT
MIRROR
21
Q3OP
V TO I
RF3N 10
11
12
13
14
VPOS
SDO
RF4P
RF4N
15
16
COMM VPOS
17
18
19
20
LODC
I4OP
Q4OP
VNEG
06587-051
AD8339
Figure 51. AD8339 Block Diagram
The AD8339 is a quad I/Q demodulator with a programmable
phase shifter for each channel. The primary application is
phased array beamforming in medical ultrasound. Other
potential applications include phased array radar and smart
antennas for mobile communications. The AD8339 can also be
used in applications that require multiple well matched I/Q
demodulators. The AD8339 is architecturally very similar to its
predecessor, the AD8333. The major differences are
•
The addition of a serial (SPI) interface that allows daisy
chaining of multiple devices
•
Reduced power per channel
Figure 51 shows the block diagram and pinout of the AD8339.
Four RF inputs that accept signals from the RF sources and a
local oscillator (applied to differential input pins marked 4LOP
and 4LON) common to all channels comprise the analog inputs.
Each channel has the option to program 16 delay states/360° (or
22.5°/step) selectable via the SPI port. The part has two reset
inputs: RSET is used to synchronize the LO dividers in multiple
AD8339s used in arrays; RSTS is used to set the SPI port bits to
all zeros. This can be useful in testing or to quickly turn off the
device without first programming the SPI port.
Each of the current formatted I and Q outputs sum together for
beamforming applications. Multiple channels are summed and
converted to a voltage using a transimpedance amplifier. If
desired, channels can also be used individually.
Rev. 0 | Page 18 of 36
AD8339
QUADRATURE GENERATION
The internal 0° and 90° LO phases are digitally generated by a
divide-by-four logic circuit. The divider is dc-coupled and
inherently broadband; the maximum LO frequency is limited
only by its switching speed. The duty cycle of the quadrature LO
signals is intrinsically 50% and is unaffected by the asymmetry
of the externally connected 4LO input. Furthermore, the divider
is implemented such that the 4LO signal reclocks the final flipflops that generate the internal LO signals and thereby minimizes
noise introduced by the divide circuitry.
For optimum performance, the 4LO input is driven differentially,
but can also be driven single-ended. A good choice for a drive is
an LVDS device as is done on the evaluation board. The commonmode range on each pin is approximately 0.2 V to 3.8 V with
the nominal ±5 V supplies.
The minimum 4LO level is frequency dependent when driven
by a sine wave. For optimum noise performance, it is important
to ensure that the LO source has very low phase noise (jitter)
and adequate input level to ensure stable mixer-core switching.
The gain through the divider determines the LO signal level vs.
RF frequency. The AD8339 can be operated to very low frequencies at the LO inputs if a square wave is used to drive the LO, as
is done with the LVDS driver on the evaluation board.
Following the phase shift circuitry, the differential current
signal is converted from differential to single-ended via a
current mirror. An external transimpedance amplifier is needed
to convert the I and Q outputs to voltages.
Table 4. Phase Select Code for Channel-to-Channel Phase Shift
Φ Shift
0°
22.5°
45°
67.5°
90°
112.5°
135°
157.5°
180°
202.5°
225°
247.5°
270°
292.5°
315°
337.5°
PHx3 (MSB)
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
PHx2
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
PHx1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
PHx0 (LSB)
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
DYNAMIC RANGE AND NOISE
www.BDTIC.com/ADI
Beamforming applications require a precise channel-to-channel
phase relationship for coherence among multiple channels. A
reset pin is provided to synchronize the LO divider circuits in
different AD8339s when they are used in arrays. The RSET pin
resets the dividers to a known state after power is applied to
multiple AD8339s. A logic input must be provided to the RSET
pin when using more than one AD8339. Note that at least one
channel must be enabled for the LO interface to also be enabled
and the LO reset to work. See the Reset Input section in the
applications section for more detail.
I/Q DEMODULATOR AND PHASE SHIFTER
The I/Q demodulators consist of double-balanced Gilbert cell
mixers. The RF input signals are converted into currents by
transconductance stages that have a maximum differential input
signal capability of 2.8 V p-p. These currents are then presented
to the mixers, which convert them to baseband (RF − LO) and
twice RF (RF + LO). The signals are phase shifted according to
the codes programmed into the SPI latch (see Table 4); the
phase bits are labeled PHx0 through PHx3 where 0 indicates
LSB and 3 indicates MSB. The phase shift function is an integral
part of the overall circuit (patent pending). The phase shift
listed in Column 1 of Table 4 is defined as being between the
baseband I or Q channel outputs. As an example, for a common
signal applied to a pair of RF inputs to an AD8339, the baseband
outputs are in phase for matching phase codes. However, if the
phase code for Channel 1 is 0000 and that of Channel 2 is 0001,
then Channel 2 leads Channel 1 by 22.5°.
Figure 52 is an interconnection block diagram of all four channels
of the AD8339. More channels are easily added to the summation
(up to 16 when using an AD8021 as the summation amplifier)
by wire-OR connecting the outputs as shown for four channels.
For optimum system noise performance, the RF input signal is
provided by a very low noise amplifier, such as the LNA of the
AD8332/AD8334 or the AD8335. In beamforming applications,
the I and Q outputs of a number of receiver channels are summed
(for example, the four channels illustrated in Figure 52). The
dynamic range of the system increases by the factor 10log10(N),
where N is the number of channels (assuming random uncorrelated noise). The noise in the 4-channel example of Figure 52 is
increased by 6 dB while the signal quadruples (+12 dB), yielding
an aggregate SNR improvement of (+12 − 6) = +6 dB.
Judicious selection of the RF amplifier ensures the least degradation in dynamic range. The input referred spectral voltage noise
density (en) of the AD8339 is nominally ~11 nV/√Hz. For the
noise of the AD8339 to degrade the system noise figure (NF) by
1 dB, the combined noise of the source and the LNA should be
approximately twice that of the AD8339, or 22 nV/√Hz. If the
noise of the circuitry before the AD8339 is less than 22 nV/√Hz,
the system NF degrades more than 1 dB. For example, if the
noise contribution of the LNA and source is equal to the AD8339,
or 11 nV/√Hz, the degradation is 3 dB. If the circuit noise
preceding the AD8339 is 1.3× as large as that of the AD8339 (or
~14 nV/√Hz), the degradation is 2 dB. For a circuit noise 1.45×
that of the AD8339 (16 nV/√Hz), the degradation is 1.5 dB.
Rev. 0 | Page 19 of 36
AD8339
To determine the input referred noise, it is important to know
the active low-pass filter (LPF) values RFILT and CFILT, shown in
Figure 52. Typical filter values for a single channel are 1.58 kΩ
and 1 nF, and implement a 100 kHz single-pole LPF. In the case
that two channels are summed, as is done on the evaluation
board, the values are the same as for a single channel of the
AD8333, namely 787 Ω and 2.2 nF.
frequency, thereby increasing the gain. The factor limiting the
magnitude of the gain is the output swing and drive capability
of the op amp selected for the I-to-V converter, in this instance,
the AD8021.
If the RF and LO are offset by 10 kHz, the demodulated signal is
10 kHz and is passed by the LPF. The single-channel mixing gain,
from the RF input to the AD8021 output (for example, I1´, Q1´)
is approximately 1.7 (4.7 dB) for 1.58 kΩ and 1 nF, or 6 dB less
when a single channel is operated on the evaluation board
(×0.85; −1.3 dB). This, together with the 11 nV/√Hz of AD8339
noise, results in ~18.7 nV/√Hz at the AD8021 output. Because
the AD8021, including the 1.58 kΩ feedback resistor,
contributes another 6.3 nV/√Hz, the total output referred noise
is approximately 19.7 nV/√Hz. This value can be adjusted by
increasing the filter resistor while maintaining the corner
SUMMATION OF MULTIPLE CHANNELS
(ANALOG BEAMFORMING)
AD8332, AD8334 LNA
OR AD8335 PREAMP
Beamforming, as applied to medical ultrasound, is defined as the
phase alignment and summation of signals generated from a
common source, but received at different times by a multielement ultrasound transducer. Beamforming has two functions:
it imparts directivity to the transducer, enhancing its gain and it
defines a focal point within the body from which the location of
the returning echo is derived. The primary application for the
AD8339 is in analog beamforming circuits for ultrasound.
AD8339
RFB
TRANSMITTER
Because any amplifier has limited drive capability, there is a
finite number of channels that can be summed. This is explained
in detail in the Channel Summing section.
2
2
2
2
I1
Φ
2
T/R
SW
Q1
Φ
www.BDTIC.com/ADI
TRANSDUCER
RFB
2
2
CFILT
I2
Φ
*
2
T/R
SW
2
2
RFILT
ΣI
ADC 16-BIT
570kSPS
I DATA
Q2
Φ
AD8021
RFB
2
I3
Φ
CFILT
AD7665 OR
AD7686
2
RFB
2
2
2
2
2
2
Q3
Φ
I4
Φ
*
RFILT
ΣQ
ADC 16-BIT
570kSPS
Q DATA
AD8021
2
0° 90°
QUADRATURE
DIVIDER
Q4
Φ
SDI
CONTROLLER
CLOCK
DATA
SYSTEM TIMING
Figure 52. Interconnection Block Diagram for AD8339
Rev. 0 | Page 20 of 36
06587-052
T/R
SW
T/R
SW
2
AD8339
The AD8339 integrates the phase shifter, frequency conversion,
and I/Q demodulation into a single package, and directly yields
the baseband signal. Figure 53 is a simplified diagram showing
the idea for all four channels. The ultrasound wave (USW) is
received by four transducer elements, TE1 through TE4, in an
ultrasound probe and generates signals E1 through E4. In this
example, the phase at TE1 leads the phase at TE2 by 45°.
PHASE COMPENSATION AND ANALOG
BEAMFORMING
Modern ultrasound machines used for medical applications
employ an array of receivers for beamforming, with typical CW
Doppler array sizes of up to 64 receiver channels that are phase
shifted and summed together to extract coherent information.
When used in multiples, the desired signals from each of the
channels can be summed to yield a larger signal (increased by a
factor N, where N is the number of channels), while the noise is
increased by the square root of the number of channels. This
technique enhances the signal-to-noise performance of the
machine. The critical elements in a beamformer design are the
means to align the incoming signals in the time domain, and
the means to sum the individual signals into a composite whole.
CHANNEL SUMMING
The circuit shown in Figure 55 of the AD8333 data sheet can be
used to sum 32 channels of the AD8339. The reason for this is
that the peak output currents of the AD8339 are approximately
half of those the AD8333. As is explained in the Channel Summing section of the AD8333 data sheet, the channel-summing
limit relates directly to the current drive capability of the amplifier
used to implement the active low-pass filter and current-tovoltage converter shown in Figure 52. The maximum sum,
when the AD8021 is used, is 16 channels of the AD8339 vs.
eight channels of the AD8333; that is, four AD8339s (4 × 4 =
16 channels) can be summed in one AD8021.
In traditional analog beamformers incorporating Doppler, a
V-to-I converter per channel and a crosspoint switch precede
passive delay lines used as a combined phase shifter and
summing circuit. The system operates at the carrier frequency
(RF) through the delay line, which also sums the signals from
the various channels, and then the combined signal is downconverted by a very large dynamic range I/Q demodulator.
In a real application, the phase difference depends on the
element spacing, λ (wavelength), speed of sound, angle of
incidence, and other factors. The signals E1 through E4 are
amplified 19 dB by the low noise amplifiers in the AD8334;
for lower performance portable ultrasound applications, the
combination of the AD8335 and the AD8339 result in the
lowest power per channel. For optimum signal-to-noise
performance, the output of the LNA is applied directly to the
input of the AD8339. In order to sum the signals E1 through
E4, E2 is shifted 45° relative to E1 by setting the phase code in
Channel 2 to 0010, E3 is shifted 90° (0100), and E4 is shifted
135° (0110). The phase aligned current signals at the output of
the AD8339 are summed in an I-to-V converter to provide the
combined output signal with a theoretical improvement in
dynamic range of 6 dB for the four channels.
The resultant I and Q signals are filtered and then sampled by
two high resolution analog-to-digital converters. The sampled
signals are processed to extract the relevant Doppler information.
www.BDTIC.com/ADI
Alternatively, the RF signal can be processed by downconversion on each channel individually, phase shifting the
downconverted signal, and then combining all channels. The
AD8333 and the AD8339 implement this architecture. The
downconversion is done by an I/Q demodulator on each channel,
and the summed current output is the same as in the delay line
approach. The subsequent filters after the I-to-V conversion
and the ADCs are similar.
TRANSDUCER
ELEMENTS TE1
THROUGH TE4
CONVERT US TO
ELECTRICAL
SIGNALS
AD8332
CH 1
PHASE SET
FOR 135°
LAG
S1
19dB
LNA
19dB
LNA
CH 2
PHASE SET
FOR 90°
LAG
S2
E2
CH 3
PHASE SET
FOR 45°
LAG
S3
19dB
LNA
CH 4
PHASE SET
FOR 0°
LAG
S4
19dB
LNA
45°
E3
135°
SUMMED
OUTPUT
S1 + S2 + S3 + S4
E4
Figure 53. Simplified Example of the AD8339 Phase Shifter
Rev. 0 | Page 21 of 36
06587-053
90°
S1 THROUGH S4
ARE NOW IN
PHASE
E1
0°
4 US WAVES
ARE DELAYED
45° EACH WITH
RESPECT TO
EACH OTHER
AD8339
PHASE BIT
SETTINGS
AD8339
SERIAL INTERFACE
The AD8339 contains a 4-wire SPI-compatible digital interface
(SDI, SCLK, CSB, and SDO). The interface comprises a 20-bit
shift register plus a latch. The shift register is loaded MSB first.
Phase selection and channel enabling information are contained
in the 20-bit word. Figure 54 is a bit map of the data-word, and
Figure 55 is the timing diagram.
ENBL BITS
When all four ENBL bits are low, only the SPI port is powered
up. This feature allows for low power consumption (~13 mW
per AD8339 or 3.25 mW per channel) when the CW Doppler
function is not needed. Because the SPI port stays alive even
with the rest of the chip powered down, the part can be awakened
again by simply programming the port. As soon as the CSB
signal goes high, the part turns on again. Note that this takes a
fair amount of time because of the external capacitor on the
LODC pin. It takes ~10 μs to 20 μs with the recommended 0.1 μF
decoupling capacitor. The decoupling capacitor on this pin is
intended to reduce bias noise contribution in the LO divider
chain. The user can experiment with the value of this decoupling
capacitor to determine the smallest value without degrading the
dynamic range within the frequency band of interest.
The shift direction is to the right with MSB first. Because the
latch is implemented with D-flip-flops (DFF) and CSB acts as
the clock to the latch, anytime that CSB is low, the latch flipflops monitor the shift register outputs. As soon as CSB goes
high, the data present in the register is latched. New data can be
loaded into the shift register at any time.
Twenty bits are required to program each AD8339 and the data
transfers from the register to the latch when CSB goes high.
Depending on the data, the corresponding channels are
enabled, and the phases are selected. Figure 55 illustrates the
timing for two sequentially programmed devices.
The SPI also has an additional pin that can be used in a test
mode, or as a quick way to reset the SPI and depower the chip.
All bits in both the shift register and the latch are reset low
when Pin RSTS is pulled above ~1.5 V. For quick testing
without the need to program the SPI, the voltage on the RSTS
pin should be first pulled high and then pulled to −1.4 V; this
enables all four channels in the (I = 1, Q = 0) state (all phase bits
are 0000); the channel enable bits are all set to 1.
Note that the data is latched into the register flip-flops on the
rising edge of SCLK, which means that the data output
transitions when SDO goes high.
www.BDTIC.com/ADI
TO PHASE SELECT AND
BIAS BLOCKS FOR
TO CHANNEL 1 PHASE TO CHANNEL 2 PHASE TO CHANNEL 3 PHASE TO CHANNEL 4 PHASE
CHANNEL ENABLES
SELECT BLOCK
SELECT BLOCK
SELECT BLOCK
SELECT BLOCK
CSB
ENABLE BITS
PH SEL CH 1
LATCH
LSB
CH1 CH2
MSB LSB
CH3 CH4 CH1
SHIFT
REGISTER
LSB
CH1 CH2
CH3
MSB LSB
CH4 CH1
PH SEL CH 2
CH1
MSB LSB
CH1 CH1 CH2
CH1
CH1
MSB LSB
CH1 CH2
PH SEL CH 3
CH2
MSB LSB
CH2 CH2 CH3
CH2
CH2
MSB LSB
CH2 CH3
PH SEL CH 4
CH3
MSB LSB
CH3 CH3 CH4
CH4
CH4
MSB
CH4
CH3
CH3
MSB LSB
CH3 CH4
CH4
CH4
MSB
CH4
SDI
SCLK
SDO
TO NEXT
AD8339
06587-054
RSTS
06587-055
Figure 54. Serial Interface Showing the 20-Bit Shift Register and Latch
Figure 55. Timing Diagram
Rev. 0 | Page 22 of 36
AD8339
APPLICATIONS
The AD8339 is the key component of a phase shifter system
that aligns time-skewed information contained in RF signals.
Combined with a variable gain amplifier (VGA) and a low noise
amplifier (LNA) as in the AD8332/AD8334/AD8335 VGA
family, the AD8339 forms a complete analog receiver for a high
performance ultrasound CW Doppler system.
LOGIC INPUTS AND INTERFACES
The SDI, SCLK, SDO, CSB, and RSET pins are CMOS compatible to 1.8 V. The threshold of the RSTS pin is 1.5 V, with a
hysteresis of ±0.3 V. Each logic input pin is Schmitt trigger
activated, with a threshold centered at ~1.3 V, and a hysteresis
of ±0.1 V around this value.
The only logic output, SDO, generates a signal that has a logic
low level of ~0.2 V and a logic high level of ~1.9 V to allow for
easy interfacing to the next AD8339 SDI input. Note that the
capacitive loading for the SDO pin should be kept as small as
possible (< 5 pF), ideally only a short trace to the SDI pin of the
next chip. The output slew is limited to approximately ±500 μA,
which limits the speed when a large capacitor is connected.
Excessive values of parasitic capacitance on the SDO pin can
affect the timing and loading of data into the next chip’s
SDI input.
The RSET mechanism also allows the measurement of nonmixing gain from the RF input to the output. The rising edge of
the active high RSET pulse can occur at any time; however, the
duration should be ≥ 20 ns minimum. When the RSET pulse
transitions from high to low, the LO dividers are reactivated on
the next rising edge of the 4LO clock. To guarantee synchronous
operation of an array of AD8339s, the RSET pulse must go low
on all devices before the next rising edge of the 4LO clock.
Therefore, it is best to have the RSET pulse go low on the falling
edge of the 4LO clock; at the very least, the tSETUP should be
≥ 5 ns. An optimal timing setup would be for the RSET pulse to
go high on a 4LO falling edge and to go low on a 4LO falling
edge; this gives 10 ns of setup time even at a 4LO frequency of
50 MHz (12.5 MHz internal LO).
Check the synchronization of multiple AD8339s using the
following procedure:
1.
Activate at least one channel per AD8339 by setting the
appropriate channel enable bit in the serial interface.
2.
Set the phase code of all AD8339 channels to the same
logic state, for example, 0000.
3.
Apply the same test signal to all devices that generates a
sine wave in the baseband output and measure the output
of one channel per device.
www.BDTIC.com/ADI
RESET INPUT
The RSET pin is used to synchronize the LO dividers in AD8339
arrays. Because they are driven by the same internal LO, the
four channels in any AD8339 are inherently synchronous.
However, when multiple AD8339s are used, it is possible that
their dividers wake up in different phase states. The function of
the RSET pin is to phase align all the LO signals in multiple
AD8339s.
The 4LO divider of each AD8339 can initiate in one of four
possible states—0°, 90°, 180°, and 270°, relative to other
AD8339s. The internally generated I/Q signals of each AD8339
LO are always at a 90° angle relative to each other, but a phase
shift can occur during power-up between the dividers of
multiple AD8339s used in a common array.
The LO divider reset function has been improved in the
AD8339 over the AD8333. The RSET pin still provides an
asynchronous reset of the LO dividers by forcing the internal
LO to hang; however, in the AD8339, the LO reset function is
fast and does not require a shutdown of the 4LO input signal.
4.
Apply a RSET pulse to all AD8339s.
5.
Because all the phase codes of the AD8339s should be the
same, the combined signal of multiple devices should be N
times greater than a single channel. If the combined signal
is less than N times one channel, then one or more of the
LO phases of the individual AD8339s is in error.
LO INPUT
The LO input is a high speed, fully differential, analog input
that responds to differences in the input levels (and not logic
levels). The LO inputs can be driven with a low common-mode
voltage amplifier, such as the National Semiconductor DS90C401
LVDS driver. The graph in Figure 21 shows the range of
common-mode voltage. Logic families, such as TTL or CMOS,
are unsuitable for direct coupling to the LO input.
Rev. 0 | Page 23 of 36
AD8339
EVALUATION BOARD
Figure 56 is a photograph of the AD8339 evaluation board, and
the schematic diagram is shown in Figure 61, Figure 62, and
Figure 63. Four single-ended RF inputs can be phase aligned
using the LNA inputs of an AD8334 and the 16 phase adjustment options of the AD8339. The RF input signals can be
derived from three sources, user selectable by jumpers. Test
points enable signal tracing at various circuit nodes.
The AD8339 requires dual power supplies, the AD8334 and
digital section only a single supply. A 3.3 V on-board regulator
provides power for the USB and EEPROM devices. The AD8339
can be configured using the software provided on the CD-ROM
included with the board, or using an external digital pattern
generator via the 20-pin flat-cable connector.
06587-056
www.BDTIC.com/ADI
Figure 56. AD8339 Evaluation Board
Rev. 0 | Page 24 of 36
AD8339
CONNECTIONS TO THE BOARD
Table 5 is a list of equipment required to activate the board with
suggested test equipment, and Figure 59 shows a typical setup.
In order to phase align any two RF input signals, the RF and
clock inputs must be coherent, that is, from the same timing
source. Many laboratory signal generators have this capability,
including the Rohde & Schwartz model shown in Table 5. Other
signal generators can also be used; the only requirement is that
they have external clock inputs and outputs. Selecting the frequency of the generators is quite simple. As an example, select
an RF frequency of interest, for example 5 MHz. Then select the
4LO frequency, which is four times the RF frequency, in this
example 20 MHz. The output frequency is 0 Hz—note that the
AD8021 outputs are at either a positive or negative dc voltage
under this condition under perfect RF and 4LO frequency lock;
more likely the signal is slowly varying if the lock is not perfect.
To detect an output, advance or retard the RF frequency by the
desired baseband frequency. A baseband frequency of 10 kHz at
the output results from an RF frequency of 5.01 MHz or 4.99 MHz.
Table 5. Recommended Equipment List
Description
Signal Generators (2), with
Synchronizing Connectors
4-Channel Oscilloscope
Suggested Equipment
Rohde & Schwartz SMT3 or
equivalent
Tektronix DPO7104 or
equivalent
Agilent E3631A or equivalent
Tektronix P6104 or equivalent
Using a common input signal source as shown in Figure 59, the
same input is applied to all four channels of the AD8339. To
observe an output at the I or Q connectors, simply enable the
appropriate channel or channels using the menu shown in
Figure 60. For example, if only Channel 1 is enabled and the
phases set to 0°, a waveform is seen at the I1 + I2 and Q1 + Q2
outputs. If Channel 2 is enabled with the phase also set to 0°,
the amplitude of the waveforms double. If Channel 1 is 0° and
Channel 2 phase set to 180°, the output becomes zero because
the phases of the two channels cancel.
When using the common input drive mode, it is important that
only the top two positions of P4A and P4B be used to avoid
shorting the LNA outputs together.
Independent Channel Drive
Independent input mode means that each channel is driven by
an LNA. Of course, the LNA inputs of the AD8334 can be
driven by up to four independent signal generators or from a
single generator. If the user chooses this mode, it is important
not to connect the LNA inputs in parallel because of the active
matching feature. Any standard splitter can be used.
AD9271 Input Drive
Connectors P3A, P3B and P4A, P4B are configured to route
input signals from the AD8334 LNA outputs or from an
AD9271 evaluation board. The AD9271 is an octal ultrasound
front end with a 12-bit ADC for each channel. When using an
AD9271 as an input drive, consult the AD9271 data sheet for
setup details.
www.BDTIC.com/ADI
Power Supplies
Scope Probes (4)
TEST CONFIGURATIONS
The three test configuration options for the AD8339-EVALZ
are common input, independent input, and AD9271 drive.
Common Input Signal Drive
Figure 57 is a block diagram showing the simplest way to use
the evaluation board, with a common signal applied to all four
AD8339 inputs in parallel. Boards are configured this way as
shipped. The inputs of each of the channels are connected in
common by means of jumpers, as shown in Table 6, although
they can be just as easily connected to any of the AD8334 LNA
outputs. As seen in Figure 62, two pairs of summing amplifiers
provide the I and Q outputs so that Channel 1 and Channel 2
can be observed independently of Channel 3 and Channel 4.
The AD9271 board is attached to the AD8339 by inserting the
three plastic standoffs into the three guide holes in the AD8339EVALZ board; all the jumpers in P3 and P4 are removed. The
bottom connectors of the AD9271 board engage P3 and P4 and
route the LNA outputs of the AD9271 to the AD8339. Figure 58
is a photograph of the two boards attached.
Table 6. P3, P4 Input Jumper Configuration
Common Input
P4A-1 to P4B-1, top two
positions (2)
RF12N, RF12P, RF23N, RF23P,
RF34N, RF34P
Rev. 0 | Page 25 of 36
Independent Input
P3A-1 to P3B-1, P4A-1 to
P4B-1
P3A-1 to P3B-1, P4A-1 to
P4B-1, all positions (8)
AD8339
COMMON
SIGNAL
PATH
AD8334
LNA
AD8339
I1
CH1
RF
Q1
Q1 + Q2
I TO V
I1 + I2
I2
CH2
RF
Q2
I TO V
I3
CH3
RF
Q3
I TO V
I3 + I4
I4
Q4
Q3 + Q4
I TO V
06587-057
CH4
RF
Figure 57. AD8339 Test Configuration—Common Signal Input Drive
06587-058
www.BDTIC.com/ADI
Figure 58. AD8339-EVALZ with AD9271 Attached as Input Source
Rev. 0 | Page 26 of 36
AD8339
TOP:
SIGNAL GENERATOR FOR 4LO INPUT (e.g., 20MHz, 1Vp-p)
BOTTOM:
SIGNAL GENERATOR FOR RF INPUT (e.g., 5.01MHz)
POWER SUPPLY
PERSONAL
COMPUTER
SYNCHRONIZE
GENERATORS
USB
CABLE
+5V
–5V
POWER
SPLITTER
4LO
INPUT
www.BDTIC.com/ADI
06587-059
OUTPUTS
Figure 59. AD8339-EVALZ Typical Test Setup
Rev. 0 | Page 27 of 36
AD8339
Using the SPI Port
Channel and phase selection are accessed via the SPI port on
the AD8339 and the evaluation board provides two means of
access. If it is desired to exercise the SPI input with custom
waveforms, the SDI, SCLK, and CSB are available at the
auxiliary connector P1. A digital pattern generator can be
programmed in conformance with the timing diagram shown
in Figure 55.
The most convenient way to select channels and phase delays
is through the USB port of a PC using the executable program
provided on the CD-ROM or the Analog Devices, Inc. website.
Copy the .EXE and .MSI files into the same folder on the PC.
Double-click on the .EXE file and the program will self-install
and place a shortcut on the desktop. Double-clicking on the
desktop icon brings up the control menu, as shown in Figure 60.
The menu consists of an array of radio buttons that are self-
explanatory. Channels are enabled or disabled by clicking on the
boxes in the list, and the 16 phase options are selected from a
drop-down menu for each of the channels.
Hardwired Jumpers
Hardwired jumpers provide for interconnection of channels
and as a means for measuring output voltages at various
strategic modes.
When shipped, the board is configured to connect all the
AD8339 RF inputs to a single LNA output. In this configuration,
the phases of the four channels can be shifted throughout the
full range and the outputs viewed on a multichannel scope
using one of the channels as a reference. To operate all the LNA
channels independently, it is only necessary to move the input
shorting jumpers to the channel RF outputs.
06587-060
www.BDTIC.com/ADI
Figure 60. SPI Software Control Menu
Table 7. Jumper and Header List
Jumper, Header
AxSHT
CSB
CSBG
EN12, EN34
G12, G34
I1234
Q1234
RF1 to RF4
RSTS
RSET
SDIG
SCLK
SDI
SLKG
VO1 to VO4
4LO
Description
Shorts the current summing outputs—shipped omitted
Connects the chip select input to the connector or the USB inputs—normally connected to USB (test)
Grounds the CSB input—shipped omitted
Enables or disables Channel 1 through Channel 4—boards shipped enabled
Connects gain pin for Channel 1 through Channel 4 to ground—boards are shipped with these jumpers inserted
Sums all four I-channel current outputs together—shipped omitted
Sums all four Q-channel current outputs together—shipped omitted
Test points for the LNA outputs—a differential probe fits these
Resets the SPI input—shipped omitted
Resets the local oscillator input—shipped omitted
Grounds the SDI input
Connects the S-clock input to the connector or the USB inputs—normally connected to USB (test)
Connects the serial data input to the connector or the USB input—normally connected to USB (test)
Grounds the S-clock input—shipped omitted
Test points for the VGA outputs—a differential probe fits these
Test pins for the 4LO level shifter output—a differential probe fits these
Rev. 0 | Page 28 of 36
AD8339
LOP1 LON1
RF1
C67
0.1μF
C43
0.1μF
R44
20Ω
LON2
R43
20Ω
RF2
LOP2
5
C44
0.1μF
C45
0.1μF
5V
8
L17
120nH
C87
0.1μF
C88
0.1μF
9
VIN2
VPS 2
U1
AD8334
VPS 3
49
50
VCM1
51
EN34
52
EN12
53
54
GAIN12
LOP2
VCM2
COM12
48
VOH1
47
NC
VOL 1
46
NC
VPS12
45
VOL 2
44
NC
VOH2
43
NC
COM12
42
MODE
41
NC
40
VIP3
VOH3
38
NC
12
LOP3
VOL 3
37
13
L14
120 nH
5V
C65
0.1μF
C55
0.1μF
L8
120nH
L9
120 nH
C62
0.1μF
IN4S
C12
22pF
CFB4
18nF
R47
20Ω
R48
20Ω
L12
120nH
C63
0.1μF
C56
0.1μF
C64
0.1μF
5V
NC
33
32
31
30
29
28
27
VPS4
26
VIP4
VIN4
25
R64
0Ω
C49
0.1μF
C50
0.1μF
RFB4
274Ω
IN4
24
23
LON4
22
21
IN3S
VCM3
INH3
VCM4
16
HILO
34
NC
CLMP34
VOH4
GAIN34
LMD3
LOP4
15
COM4X
35
NC
LMD4
VOL 4
COM34
L15
120 nH
36
COM3X
C10
22pF
C61
0.1μF
39
14
17 COM3
C48
0.1μF
VPS34
LON3
20
RFB3
274Ω
CFB3
18nF
11
INH4
LON3
COM34
19
R45
20Ω
VIN3
COM4
C47
0.1μF
10
18
R46
20Ω
RF3
IN3
55
58
59
60
61
62
63
PIN 1
IDENTIFIER
C59
0. 1μF
www.BDTIC.com/ADI
C46
0. 1μF
LOP3
C58
0.1μF
6 VIP2
7
L16
120 nH
VPS1
LON2
DIS
R63
0Ω
VIN1 56
4
LOP1
COM2X
LO N1
3
COM1X
LMD2
INH1
2
EN34
DIS
C57
0. 1μF
C85
0.1μF
RF4
5V
LON4 LOP4
Figure 61. Schematic—LNA Section
Rev. 0 | Page 29 of 36
06587-061
RFB2
274Ω
EN
EN12
C66
0.1μF
CLMP12
CFB2
18nF
COM2
INH2
COM1
64
C60
0.1μF
1
L13
120nH
C53
0.1μF
C54
0.1μF
C68
0.1μF
C8
22pF
5V
EN
C86
0. 1μF
C6
22pF
L7
IN2 120 nH
IN2S
RFB1
274 Ω
57
IN1S
CFB1
18nF
R49
20Ω
VIP1
L10
120nH
LMD1
IN1
5V
R50
20Ω
AD8339
5V
R3
1kΩ
R4
1kΩ
P1
CSB
SCLK
(SHT3)
SDI
RSTS
R32
2.8kΩ
R61
0Ω
VPIS
R62
0Ω
R59
0Ω
2
R72
2.8kΩ
VPOS
28
VPOS
27
DUT
AD8339
SCLK
-
1
I 1234
R14
0Ω
6
I1+I2
5
+
3
Q1234
C81
2.2nF
7
AD8021
8
R18
0Ω
C82
5PF
4
-VA
C24
0.1µF
VPOS
C30
0.1µF
U7
DSC90C401
RF3P
I3OP
22
R8
0Ω
RF3N
Q3OP
21
R7
0Ω
R28
C28
3.48kΩ 0.1µF
RF34P
RF34N
SDO
R34
2.8kΩ
N23
P12
P23
R51
0Ω
I4O P
2
8
R6
0Ω
3
I4OP
-
C80
5PF
4
C22
0. 1µF
5V
VNEG
R42
787 Ω
7
1 U6
C32
2.2nF
2
Q4OP
8
3
AD8021
5
+
4
R38
Q3+Q4
0Ω
6
C33
5PF
-VA
C52
0. 1µF
2
1
LON4
LOP4
1
LON3
-VA
C51
0.1µF
R2
0Ω
I3+I 4
5
+
R5
0Ω
C20
0.1µF
R54
49.9Ω
R11
0Ω
6
AD8021
LOP
C79
2.2nF
7
P34
LOP3
LON2
LOP2
LON1
C21
0. 1µF
R9
0Ω
1
COMPONENTS SHOWN IN
GRAY ARE NOT
INSTALLED
2
1
1
P4A
P4B
VA
VA
R36
5.23kΩ
R30
4.22kΩ
N34
R10
787 Ω
C18
0.1µF
VPOS
W7
W8
RF3P
N12
SDO R56
0Ω
W6
SDO
RF2N
W11
RF2P
W10
C17
0.1µF
RF1N
W13
R33
2. 8kΩ
5V
W9
RF3N
VPOS
R37
1.5kΩ
VNEG
C29
0. 1µF
C19
0.1µF
R65
4.22kΩ
4
2
17
12
11
VPOS
VPOS
R57
0Ω
VNEG
23
Q4OP
VNEG
LODC
VPOS
COMM
10
24
15
9
VNEG
14 RF4N
8
R69
2.8kΩ
R27
100 Ω
4LO
VPOS
13 RF4P
R58
0Ω
26
25
SDO
7
4LOP
4LON
16
RF23N
RF23P
SLKG 5
R70
2.8kΩ
RF1P
W12
2
R19
0Ω
29
R13
787 Ω
C23
0.1µF
VA
R12
0Ω
Q1+Q 2
-VA
www.BDTIC.com/ADI
VPOS
RF3N
COMM
CSBG 6
CSB
C16
0.1µF
5V
R67
5.23kΩ
LOP1
R21
0Ω
COMM
RF3P
C26
0. 1µF
31
34
RSET
35
36
37
38
I2OP
C84
5PF
4
Q1OP
30
R17
0Ω
6
5
+
3
Q2OP
7
AD8021
8
VNIS
3
4
-
1
VPOS
R66
4.22kΩ
2
-5V
PIN 1
IDENTIFIER
RF2P
C83
C25
0. 1µF 2.2nF
R15
0Ω
L1
120 nH
C27 R20
0. 1µF 0Ω
CO MM
RF2N
RF1N
1
RF1P
RSTS
R71
2.8kΩ
RF2P
39
RF12P
40
RF2N
19
VNEG
VPOS
RF12N
R60
0Ω
VA
I1OP
C1
0.1µF
SDI
R68
5.23kΩ
15
17
C31
0.1µF
RSET
VPOS
5V
16
18
20
R16
787 Ω
L2
120 nH 5V
RF1P RF1N
R29
4.22kΩ
9
11
13
VNEG
R31
2.8kΩ
W1
SDI
(SHT3)
10
12
14
32
R35
5.23kΩ
W2
Q1OP
5V
SDO
3
5
7
33
SLKG
W4
CSB
(SHT3)
SCLK
1
4
6
8
I1OP
CSBG
2
FROM AD9271
P3A
P3B
C34
0.1 µF
5V
U7
1 DSC90C401
7
8
FROM LNA
OUTPUTS
6
5
Figure 62. Schematic—IQ Demodulator and Phase Shifter
Rev. 0 | Page 30 of 36
06587-062
R1
1kΩ
AD8339
5VS
-5VS
RED
VAS
GRN
-VAS
ORG
BLUE
PLUS
5V
MINUS
-VAS GND1 GND2 GND3 GND4 GND5 GND6 GND7 GND8
VAS
C9
1 µF L11
10V 120 nH
W3
+
3
2
1
OUT GND
IN
-5VS
5VS
C77
0.1 µF
A6
ADP3339AKC-3.3
OUT
TAB
C14 +
10µF
25V
L6
120 nH
C13
10µF
25V +
L5
120 nH
C11 +
10µF
25V
L4
120 nH
C15
10µF
25V+
L15
120 nH
C38
0. 1µF
C37
0. 1µF
C36
0. 1µF
C35
0. 1µF
3.3V
5V
3.3V
VA
-5V
BLK
TEST
LOOP
(8)
-VA
R41
100kΩ C5
22pF
C76
0.1 µF
C7
0.1 µF
NC
Y1
24MHz
3.3V
2
3
C69
0.1 µF
4
5
C3
12 pF
C2
12 pF
AVCC
43
3. 3V
VCC
WAKEUP
PD0/FD8
PD1/FD9
PD2/ FD10
PD3/ FD11
PD4/FD12
PD5/FD13
PD6/FD14
GND
VCC
RDY1/SLWR
NC
NC
NC
NC
48
47
46
45
44
RESET #
GND
PA7/FLAGD/SLCS
PA6/PKTEND
XTALOUT
PA5/ FIFOADR1
XTALIN
42
R40
100kΩ
41
C4
22pF
40
39
38
CSB(SHT2)
SCLK(SHT2)
SDI (SHT2)
www.BDTIC.com/ADI
C72
0.1 µF
Z1
24LC00/ P
1 A0
2 A1
VCC
R52
22. 1kΩ
R53
22. 1kΩ
8
WP 7
3 A2 SCL 6
4 VSS SDA 5
18
19
20
21
22
23
24
25
NC
17
VCC
GND
26
27
NC
36
NC
35
NC
34
NC
33
NC
32
31
NC
3.3V
C75
0.1 µF
30
NC
29
NC
28
C74
0.1 µF
C73
0.1 µF
R22
0Ω
3.3V
3.3V
16
NC
15
37
GND
CTL0/FLAGA
PB7/FD7
RESERVED
PB6/FD6
CTL1/FLAGB
PB5/FD5
IFCLK/PE0/TOUT
SCL
R39
10kΩ
CTL2/FLAGC
NC
14
GND
PB4/FD4
13
NC
VCC
NC
12
PA0/ INT0#
VCC
PB3/FD3
C71
0.1 µF
GND
NC
11
PB2/FD2
CR1
3. 3V
NC
555
499Ω
10
PA1/ INT1#
PB1/FD1
1
5V
PA2/SLOE
DMINUS
NC
VBUS
9
DPLUS
PB0/FD0
D-
PA3/WU2
U2
CY7C6801 3A-56LFXC
NC
8
AVCC
VCC
GND
4
7
C70
0.1 µF
PA4/ FIFOADR0
GND
3.3V
2
6
SDA
3.3V
A7
USB TYPE B D+
3
RDY0/SLRD
NC NC
NC NC
52
51
50
49
PD7/FD15
1
CLKOUT/PE1/T1OUT
NC
NC
54
53
55
GND
56
R23
0Ω
R24
0Ω
R25
0Ω
R26
0Ω
Figure 63. Schematic—USB
Rev. 0 | Page 31 of 36
06587-063
C78
0.1 µF
AD8339
AD8339-EVALZ ARTWORK
Figure 64 through Figure 67 show the artwork for the AD8339-EVALZ.
06587-064
www.BDTIC.com/ADI
06587-065
Figure 64. AD8339-EVALZ Component Side Copper
Figure 65. AD8339-EVALZ Wiring Side Copper
Rev. 0 | Page 32 of 36
06587-066
AD8339
Figure 66. AD8339-EVALZ Component Side Silkscreen
06587-067
www.BDTIC.com/ADI
Figure 67. AD8339-EVALZ Assembly
Rev. 0 | Page 33 of 36
AD8339
Table 8. Bill of Materials
Qty
1
1
24
Name
Test Loop
Test Loop
Header
Description
Green
Blue
Berg 2
1
1
Red
3.3 V regulator
1
Test Loop
Integrated
Circuit
Connector
63
Capacitor
2
6
1
4
Capacitor
Capacitor
Capacitor
Capacitor
4
4
4
Capacitor
Capacitor
Capacitor
1
12
LED
Test Loop
1
7
9
Integrated
Circuit
Header
Test Loop
Berg 3
Black
9
Connector
SMA right angle
17
Bead
Ferrite, 120 μH
1
3
Connector
Connector
2
Connector
2
3
31
Connector
Resistor
Resistor
20-pin RT/A latch eject
Header, vertical, 1" 1X4
30AU
Header, vertical, 1" 3X4P
30AU
Header, 100 dual str, 2 × 4
1.0 kΩ, 1/16 W, 1%, 0603
0 Ω, 5%, 1/10 W, 0603
4
1
1
1
Resistor
Resistor
Resistor
Resistor
787 Ω, 1%, 1/16 W, 0603
100 Ω, 1%, 1/16 W, 0603
3.48 kΩ, 1%, 1/16 W, 0603
1.5 kΩ, 1%, 1/16 W, 0603
Right angle USB, Type B,
4-position
0.1 μF, 16 V, 0603 X7R
12 pF, 50 V, 5%, 0603
22 pF, 50 V, 5%, 0603
1 μF, 6.3 V, 10%, 0603
Tantalum 10 μF, 10 V,
20%, 3216, SMD
2.2 nF, 50 V, 10%, 0603
5 pF, 50 V, 0603
0.018 μF, 10%, 50 V, X7R,
0603
Green, USS type, 0603
Purple
Reference Designator
−5VS
−VAS
4LO, I1234, IN1S, IN2S, IN3S, IN4S, MINUS,
PLUS, Q1234, RF1, RF2, RF3, RF4, RF12N,
RF12P, RF23N, RF23P, RF34N, RF34P, VNIS,
VPIS, WQ3, W18, W19
+VS
A6
Manufacturer
Components Corp.
Components Corp.
Molex
Part Number
TP-104-01-05
TP-104-01-06
22-10-2021
Components Corp.
Analog Devices
TP-104-01-02
AD3339AKCZ-3.3-RL
A7
Tyco
292304-1
C1, C7, C16, C17, C18, C19, C20, C21, C22,
C23, C24, C25, C26, C27, C28, C29, C30,
C31, C34, C35, C36, C37, C38, C43, C44,
C45, C46, C47, C48, C49, C50, C51, C52,
C53, C54, C55, C56, C57, C58, C59, C60,
C61, C62, C63, C64, C65, C66, C67, C68,
C69, C70, C71, C72, C73, C74, C75, C76,
C77, C78, C85, C86, C87, C88
C2, C3
C4, C5, C6, C8, C10, C12
C9
C11, C13, C14, C15
Kemet
C0603C104K4RACTU
AVX
Panasonic
Panasonic
Nichicon
06035A120JAT2A
ECJ-1VC1H220J
ECJ-1VB0J105K
F931A106MAA
www.BDTIC.com/ADI
Quad I/Q demodulator
C32, C79, C81, C83
C33, C80, C82, C84
CFB1, CFB2, CFB3, CFB4
Panasonic
Panasonic
AVX
ECJ-1VB1H222K
ECJ-1VC1H050C
06035C183KAT2A
CR1
CSB, CSB-TP, I1OP, I4OP, Q1OP, Q4OP,
RESET, RSTS-TP, SCLK-TP, SDI, SDI-TP,
SDO-TP
U8
Panasonic
Components Corp.
LNJ314G8TRA
TP-104-01-07
Analog Devices
AD8339ACPZ
EN12, EN34, SDO, W1, W2, W4, W5
GND1, GND2, GND3, GND4, GND5, GND6,
GND7, GND8, GND9
I1 + I2, I3 + I4, IN1, IN2, IN3, IN4, LOP,
Q1 + Q2, Q3 + Q4
L1, L2, L3, L4, L5, L6, L7, L8, L9, L10, L11,
L12, L13, L14, L15, L16, L17
P1
P2, Positions W14 to W15, Positions W16
to W17
Positions 3A to 3B, Positions 4A to 4B
Molex
Components Corp.
22-10-2031
TP-104-01-00
Amphenol
901-143-6RFX
Murata
BLM18BA750SN1D
3M
Molex
3428-5603
22-10-2041
Samtec
TSW-104-14-G-T
Positions W10 to W13, Positions W6 to W9
R1, R3, R4
R2, R5, R6, R7, R8, R9, R11, R12, R14, R15,
R17, R18, R19, R20, R21, R22, R23, R24,
R25, R26, R38, R51, R56, R57, R58, R59,
R60, R61, R62, R63, R64
R10, R13, R16, R42
R27
R28
R37
Sullins
Panasonic
Panasonic
S2011E-04-ND
ERJ-3EKF1001V
ERJ-2GE0R00X
Panasonic
Panasonic
Panasonic
Panasonic
ERJ-3EKF7870V
ERJ-3EKF1000V
ERJ-3EKF3.48K
ERJ-3EKF1501V
Rev. 0 | Page 34 of 36
AD8339
Qty
1
2
8
2
1
1
4
1
1
4
1
1
1
1
Name
Resistor
Resistor
Resistor
Resistor
Resistor
Resistor
Resistor
Integrated
Circuit
Integrated
Circuit
Integrated
Circuit
Integrated
Circuit
Test Loop
Crystal
Integrated
Circuit
Description
10 kΩ, 1%, 1/16 W, 0603
100 kΩ, 1%, 1/16 W, 0603
20.0 Ω, 1%, 1/16 W, 0603
22.1 kΩ, 1%, 1/16 W, 0603
49.9 Ω, 1%, 1/16 W
499 Ω, 1%, 1/16 W, 0603
274 Ω, 1%, 1/16 W, 0603
Quad VGA
Reference Designator
R39
R40, R41
R43, R44, R45, R46, R47, R48, R49, R50
R52, R53
R54
R55
RFB1, RFB2, RFB3, RFB4
U1
Manufacturer
Panasonic
Panasonic
Panasonic
Panasonic
Panasonic
Panasonic
Panasonic
Analog Devices
Part Number
ERJ-3EKF1002V
ERJ-3EKF1003V
ERJ-3EKF20R0V
ERJ-3EKF2212V
ERJ-3EKF49R9V
ERJ-3EKF4990V
ERJ-3EKF2740V
AD8334ACPZ-REEL7
MCU USB peripheral,
high speed, 56 QFN
Amplifier
U2
Cypress
CY7C68013A-56LFXC
U3, U4, U5, U6
Analog Devices
AD8021ARZ
LVDS driver
U7
DS90C401M
Orange
24.00 MHz, 12 pF,
HC-49/US
EEPROM
VAS
Y1
National
Semiconductor
Components Corp.
ECS Inc.
Microchip
Technology
24LC00/P
Z1
TP-104-01-03
ECS-240-12-4X
www.BDTIC.com/ADI
Rev. 0 | Page 35 of 36
AD8339
OUTLINE DIMENSIONS
6.00
BSC SQ
0.60 MAX
0.60 MAX
PIN 1
INDICATOR
31
30
TOP
VIEW
0.50
BSC
5.75
BCS SQ
0.50
0.40
0.30
12° MAX
(BOT TOM VIEW)
10
11
21
20
0.25 MIN
4.50
REF
0.80 MAX
0.65 TYP
0.05 MAX
0.02 NOM
1.00
0.85
0.80
SEATING
PLANE
0.30
0.23
0.18
0.20 REF
4.25
4.10 SQ
3.95
EXPOSED
PAD
THE EXPOSED PAD IS NOT CONNECTED
INTERNALLY. FOR INCREASED RELIABILITY
OF THE SOLDER JOINTS AND MAXIMUM
THERMAL CAPABILITY IT IS RECOMMENDED
THAT THE PAD BE SOLDERED TO
THE GROUND PLANE.
COPLANARITY
0.08
COMPLIANT TO JEDEC STANDARDS MO-220-VJJD-2
101306-A
PIN 1
INDICATOR
40
1
Figure 68. 40-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
6 mm × 6 mm Body, Very Thin Quad
(CP-40-1)
Dimensions shown in millimeters
ORDERING GUIDE
www.BDTIC.com/ADI
Model
AD8339ACPZ 1
AD8339ACPZ-R71
AD8339ACPZ-RL1
AD8339-EVALZ1
1
Temperature Range
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
Package Description
40-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
40-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
40-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
Evaluation Board
Z = RoHS Compliant Part.
©2007 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D06587-0-8/07(0)
Rev. 0 | Page 36 of 36
Package Option
CP-40-1
CP-40-1
CP-40-1
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